1. Field of the Invention
The present invention relates to calibration of electronic devices, and more particularly, to techniques for efficiently calibrating these devices to prevent or reduce unwanted signals at the device output.
2. Description of the Related Art
It is often required that the input-output relationship of electronic components be predictable and accurate. For instance, in communication systems, signals conveying information that are transmitted and received, such as voice or data, undergo processing in several stages both before transmission and upon reception. However, several factors can contribute degradation of the signal fidelity from the time the signal is created, transmitted, received and reproduced. As a signal is processed using analog and/or digital processing (DSP) techniques, gain, offset and imbalances of phase may cause errors to be introduced into the signal, which can result in the loss of information and cause unwanted signal interferers. To exemplify how such errors are introduced and how they are compensated according to the conventional solution, a model of a generalized transmitter and receiver will be studied in more detail.
Transmitters include circuits for processing and transmitting source information over a communication channel. The source information may first be imposed on a baseband signal, and may be in the form of an analog or digital signal. Baseband signals are not usually suitable for direct transmission over the communications medium because of their relatively low frequency. Consequently, prior to transmission, baseband signals are imposed onto a carrier signal having a given carrier frequency by means of circuits called modulators. A modulator varies one or more characteristics (e.g., amplitude, phase and/or frequency) of the carrier signal to create a modified carrier signal that not only is suitable for transmission over the channel, but also contains information that enables a receiver to substantially recreate the original baseband signal.
All bandpass waveforms, such as those arising from a modulated carrier signal, may be represented in a convenient form represented by v(t)=Re{g(t)ejωct}, where Re {∘} denotes the real part of a complex number {∘}, g(t) is called the complex envelope of v(t), and fc is the associated carrier frequency, ωc=2πfc. The complex envelope g(t) may be a complex function in real time. In terms of two real functions in Cartesian coordinates, g(t)≡x(t)+jy(t), where x(t)=Re{g(t)} and y(t)=Im{g(t)}. x(t) is said to be the in-phase modulation associated with v(t), and y(t) is said to be the quadrature modulation associated with v(t). In modem communication systems, the baseband signal is often partitioned into two channels, one for x(t) called the I (in-phase) channel and one for y(t) called the Q (quadrature-phase) channel.
FIG. 1 shows a generalized transmitter that uses in-phase and quadrature-phase (I and Q) processing. As shown in FIG. 1, a modulating signal m(t) (i.e., the source baseband signal) is supplied to a baseband signal processor 1, which uses known techniques to implement a desired modulation type, for example, QPSK, BPSK, BFSK, QM, QAM, or GMSK type modulation. The baseband signal processor outputs the I and Q signals (i.e., x(t) and y(t), respectively) of the desired modulation type. The I and Q signals are supplied to respective mixer circuits 3 and 4 in RF circuitry 2. Mixer circuit 3 multiplies the I signal by a carrier signal (LO signal) supplied by a local oscillator 5 of carrier frequency fc. Mixer circuit 4 multiplies the Q signal with a 90° (or −90°) phase shifted version of the LO signal. The phase shifted signal may be supplied by inputting the LO signal into a phase shift circuit 6 (e.g., a phase splitter). The outputs of the mixers 3, 4 are combined in a summing device 7 to produce the composite signal v(t) suitable for transmission over the communication channel.
A receiver has the job of recovering the source information from the transmitted modulated carrier. In the receiver, a demodulator performs the required inverse mapping to recover the source information mapped onto the carrier signal. The most common types of receivers in use are either the heterodyne or homodyne type. A heterodyne receiver receives the input signal, amplifies it at radio frequency (RF) in a tuned stage, converts the amplified signal to the lower intermediate frequency (IF) by way of two mixers including an offset frequency LO and a 90° (or −90°) phase shifted version of the LO signal, and finally amplifies the mixed signals in tuned IF “strips” containing highly-selective passive bandpass filters. The processed IF signal is later mixed with yet another signal to generate a signal at the original baseband frequency, or is selected by way of a bandpass filter or other detecting means. By contrast, the homodyne receiver skips all processing at an intermediate frequency, and simply uses a mixer to translate the channel directly from RF to 0 Hz center frequency, and each I and Q channel is selected with a lowpass filter centered at DC (when the negative frequency axis is included).
In a modulator or a demodulator, LO-leakage at the inputs will self-mix with the LO signal (i.e., mixing the LO-leakage at the input with the same LO frequency) and lead to a zero-Hz mixing component (i.e., the DC offset) at the output. Self-mixing is mainly a concern in demodulation because LO-leakage at the demodulator input causes undesirable DC offset to appear at the demodulator output, for example, when down-converting a bandpass signal to baseband. On the other hand, LO-leakage at a modulator output often is the result of having DC offset present at the modulator input.
In an RF mixer, there will always be some amount of leakage of the LO signal. The amplitude of LO-leakage at the output will depend on how big the DC offset is at the input. In some applications, there is a need to provide DC compensation to minimize LO-leakage because LO-leakage impairs the modulation accuracy. Therefore, in these applications, the undesirable LO-leakage resulting from a DC offset must be suppressed to a minimum required value. This can be realized by adding compensating DC to the input with the sign opposite from the original DC offset. The DC compensation is generated by an adjustable signal generator whose output signal level is controlled, either directly or indirectly, by the value of a control parameter represented by a number of bits. The control parameter should be set to a value that results in a desired DC compensation level chosen to compensate for the undesirable DC offset.
In an in-phase (I) and quadrature-phase (Q) RF mixer, a number of settable bits may be provided for setting an amount of offset compensation independently in each of the I and Q channels, thus making the settings two-dimensional. For example, if seven combinations are possible for each channel, there will be a total of 49 possible combinations for setting both channels. In the production of integrated components that include adjustable DC compensation, the amount of DC compensation must be determined individually for each unit. Thus, it is desirable to make as few measurements as possible when setting compensation for LO-leakage in each of the I and Q channels. However, the number of combinations in a two-dimensional configuration leads to delays during the production of circuit components because it is often necessary to make an excessive number of measurements before determining a setting for an acceptable LO-leakage level, or before a determination can be made as to whether it is necessary to discard the circuit component entirely (e.g., when LO-leakage is greater than can be compensated for by any combination of the bits associated with each of the I and Q channels). The conventional methods for setting DC offset compensation are cumbersome, time consuming and cause excessive wear on production facilities, resulting in reduced output production and increased costs.
One solution to the problems associated with the conventional methods is to search for minimum LO-leakage by first changing the DC compensation settings on one channel while not altering the other. After finding the minimum for one channel, the other channel can be searched. By selecting which direction to search based on the difference in LO-leakage, and by starting the search on the second channel with the best setting in place on the first, the number of measurements needed is reduced and will be between 1 and 9 for the 7×7 example. For other grids, the number of measurements will be different. The worst case of 9 measurement iterations includes two “sign changes,” one for each I and Q channel, which would be necessary when a particular initial iteration direction (i.e., either in the positive or negative direction) causes the offset to increase. The number of sign changes involved on any one channel can be in the range 0 to 1, for a total number of 0 to 2 sign changes for both channels. In some instances, sign shifts may increase a level of complexity to a calibration process, but most often do not present serious problems.
The calibration methods of the related art require a number of iterative measurement steps that may exceed an acceptable level when considering factors such as increasing efficiency and speed of device production to reduce production cost and complexity. Therefore, there is a need in the art to reduce the number of measurements that would be required when calibrating these components.